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SPS Design Schematic rev2 2.pdf

Note: It is easier to refer to the schematic above when reading section 1.5.1 instead of the detailed Eagle schematic in section 1.4.1.1. This is because this section exclusively covers all SPS components. The part numbers are consistent with those in the Eagle schematic.

Front end Passive Block

Power Bus Input Choke (L201a, L201b)

Part Description:

• CMS1-11-R Common Mode Inductors 100 uH, Micro-PAC Plus Package, RoHS Compliant, Tape and Reel. http://www.cooperet.com/library/products/PM-4313%20CMS-Series.pdf.

Purpose:

• Common mode choke (balanced inductor). It is used as an EMI filter between the power bus and the SPS.

Specifications/ Calculations:

• The value was chosen through a trial and error process from the pervious LV2 SPS design. Each inductor of the choke is 100 uH. See page 13 of the CAN Node Switch Mode Power Supply (SPS) (200) section in the Component Design for LV2 Power Electronics (Except Main Battery) engineering design notes.

Power Bus Fast-Acting Fuse (F200)

Part Description:

• 500 mA, 50 V, 1206, Fast Acting Short Time Lag, RoHS Compliant, Wickmann USA Inc, FCD120500TP (Digi-Key p/n WK6213CT-ND $0.56/1) http://rocky.digikey.com/WebLib/WICKMANN/Web%20Data/FCD12.pdf.

Purpose:

• This fuse protects the SPS from currents greater than 500 mA. Its direct purpose however is to protect the power bus from a short circuit fault on the SPS side.

Specifications/ Calculations:

• Since the specified maximum SPS current is 400 mA we chose a fuse rated at 500 mA. The opening time for the fuse according to its datasheet is 1 s at a current of about 1.15 A. Currents below 1 A are several hundred seconds, therefore this fuse will only protect the SPS or power bus from gross currents due to some fault on either side (power bus or SPS) and not to keep the SPS output current within spec, that is U250's job.

DC Path From SPS Node GND to Chassis GND Resistor (R215)

Part Description:

• 100 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant, Rohm, MCR10EZHF1003 (Digi-Key p/n RHM100KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R215 provides a DC path from the SPS ground to chassis ground. See page 29 of the CAN Node Switch Mode Power Supply (SPS) (200) section in the Component Design for LV2 Power Electronics (Except Main Battery) engineering design notes.

Specifications/ Calculations:

• 100 kohm worked.

Power Bus Input Caps (C203, C204)

C203

Part Description:

• 22 uF, 25 V, Tant, T491 Series, 7343-31 (EIA), Cut Tape, RoHS Compliant???, Kemet, T491D226K025AT (Digi-Key p/n 399-3782-1-ND $0.65/1) http://www.kemet.com/kemet/web/homepage/kechome.nsf/vapubfiles/F3102T491.pdf/$file/F3102T491.pdf.

Purpose:

• C203 acts as a noise filter between the power bus and SPS. It also serves as a local energy storage node.

Specifications/ Calculations:

• It needs to have a voltage rating greater than 20 V and a low equivalent series resistance (ESR) thus a tantalum capacitor was chosen due to their low ESR at a higher capacitance.

C204

Part Description:

• 0.33 uF, 50 V, 0805, X7R, Cut Tape, RoHS Compliant, Murata Electronics North America , GRM219R71H334KA88D (Digi-Key p/n 490-3327-1-ND $3.09/10) http://search.murata.co.jp/Ceramy/image/img/PDF/ENG/GRM219R71H334KA88.pdf.

Purpose:

• C204 is a high frequency noise filter between the power bus and SPS. It did not have to have as high a capacitance as C203 so the trade off was to get a lower value at a low ESR.

Specifications/ Calculations:

• It needs to have a voltage rating greater than 20 V and the capacitance value was not very critical but should be much lower than C203.

Power Bus Input Voltage Suppressor (TVS200)

Part Description:

• 18 V, SMB, Unidirectional, Cut Tape, RoHS Non-Compliant, Diodes Inc, SMBJ18A-13 (Digi-Key p/n SMBJ18ADICT-ND $0.89/1) http://www.diodes.com/datasheets/ds19002.pdf.

Purpose:

• A transient voltage suppressor (TVS), this "zener like" diode protects the SPS (specifically U200) in the event of an overvoltage at the input.

Specifications/ Calculations:

• It should have a breakdown voltage of about 20 V (unlikely maximum bus voltage) and a current carrying capacity greater than the fuse rated current. It should have a fast response time and be unidirectional. In the event of a sustained overvoltage at the input the only allowable part which can be destroyed is the fuse, F200. That is what we want.

Circuit Breaker Block

MAX5902 Circuit-Breaker Resistor Network (R250, R251)

R250

Part Description:

• 61.9 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF6192 (Digi-Key p/n RHM61.9KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R250 is part of the UVLO resistor divider of U250.

Specifications/ Calculations:

• The UVLO voltage was specified to be 9 V. See page 8 and Figure 3 in the MAX5902 datasheet. Letting R251 = 10.0 kohm and using the typical value of V_on/!off = 1.26 V, the UVLO formula from page 8 in the datasheet is R250 = R251 * ((V_UVLO / (V_on/!off)) - 1) = 61.4 kohm. The closet standard value was 61.9 kohm.

R251

Part Description:

• 10.0 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF1002 (Digi-Key p/n RHM10.0KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R251 is part of the UVLO resistor divider of U250.

Specifications/ Calculations:

• R251 was specified to be 10.0 kohm. See page 8 and Figure 3 in the MAX5902 datasheet.

Overcurrent/Circuit-Breaker Protection (U250)

Part Description:

• MAX5902AAETT +72 V, SOT-23, Simple Swapper How-Swap Controller. Ordered samples from vendor, no Digi-Key. http://pdfserv.maxim-ic.com/en/ds/MAX5902-MAX5903.pdf.

Purpose:

• This hot-swap controller IC serves two purposes: (1) circuit-breaker and (2) the first stage of UVLO protection. The version we chose had a input voltage range of +9 V to +72 V, a 300 mV circuit-breaker threshold voltage, limited inrush current ("soft start") and was an automatic retry circuit-breaker. It also had a built-in thermal shutdown and active low power good (!PGOOD) indicator output pin. The device needed a UVLO resistor divider network (R250, R251) and an external PMOSFET (Q250) switch. There are four events which will cause Q250 to turn off: (1) if there is undervoltage at the input, (2) if there is overcurrent, (3) if the die temperature exceeds +125 C and (4) the ON/!OFF pin 6 is forced low for at least 10 ms. See the MAX5902 datasheet.

Specifications/ Calculations:

• The reasons we chose the 300 mV automatic retry circuit breaker version was that we wanted the SPS to be able to recover from a fault condition by itself and we expect that the nominal load current will not be very close to the 400 mA maximum limit but closer to 300 mA or less. Hence steady-state currents in the range of 400 mA to 500 mA qualify as an overcurrent event and should be detected. To avoid wasting power dissipated by Q250's RDS(on) and R252, those values should be kept low, therefore the voltage across them should also be low and the 300 mV threshold version satisfied that. Upon power up U250 keeps Q250 off and if trigger events (1) and (2) are non-existent, then it gradually turns Q250 on to saturation in approximately 150 ms. The drain of Q250 is gradually enhanced at a rate of about 9 V/ ms. This start sequence limits the inrush current giving some "soft-start" protection to its load. Once all transients are gone before the 150ms time period and Q250 is fully saturated, U250's circuit-breaker functionality comes up and monitors the Vds of Q250 between pins 1 and 2. Before this initial power up 150 ms period there is no circuit-breaker functionality. If any one of the 4 trigger events occurs U250 will turn Q250 off, de-assert !PGOOD (output a logic high) and reinitiate the start sequence given that the trigger event(s) disappears during the 150 ms period, if not the 150 ms period will repeat. There are two typical turn off times regarding Q250: 10 ms and 4 us. If there is an ON/!OFF or UVLO trigger event, they need to exist for 10 ms before U250 turns Q250 off, which will take an unspecified amount of time. If there is an overcurrent or temperature trigger event, then Q250 is turned off in 4 us. If the trigger events disappear after Q250 turns off within 150 ms, then the normal start sequence is reinitiated. Since the purpose of U250 was to be a circuit-breaker we decided not to use the ON/!OFF pin to turn Q250 off via any SPS feedback. Only an UVLO condition would be a trigger event. The UVLO threshold was specified as 9 V. See R250 and R251. This meant that when a trigger event other than a UVLO condition happens, U250 would turn Q250 off in 4 us and would reinitiate the start sequence after a trigger event free 150 ms time period. We needed a way for the LPC2148 (U280) microcontroller to turn off the SPS, so we connected a logic level NMOSFET (Q282) to a GPIO pin and connected the drain of Q282 to the ON/!OFF pin of U250. Also we connected the active low !PGOOD pin to another GPIO pin for monitoring and/or interrupt purposes. See the GLUE Logic section regarding U280 and Q282. See R253 and Q251 for the "overvoltage to overcurrent" trigger event emulation. One bad thing that we did not like was the relatively high supply current of U250 being 1 mA to 2 mA. We believe that U250 will draw the most current from the power bus when the SPS is in a standby/shutdown mode. NOTE: According to the MAX5902 datasheet there are two package options for the different versions of the MAX5902: a TDFN and SOT23 package. All SOT23 packages have the specification that, "This device is constructed using a unique set of packaging techniques that impose a limit on the thermal profile the device can be exposed to during board level solder attach and rework. This limit permits only the use of solder profiles recommended in the industry standard specification, JEDEC 020A, paragraph 7.6, Table 3 for IR/VPR and convection reflow. Preheating is required. Hand or wave soldering is not allowed.", which may or may not present a problem. The invoice we received for the ordered samples specified that they were MAX5902AAETT+ in a SOT23 package (they were in the SOT23 package). The datasheet however specifies that the MAX5902AAETT is a TDFN part (which does not have a special solder specification) and all SOT23 part numbers have the "*" symbol suffix. The "+" symbol suffix in the invoice means its a lead-free part. There are some discrepencies.

MAX5902 External P-MOSFET (Q250)

Part Description:

• -60 V, -3 A, SOT-23-6, P-Channel MOSFET, Cut Tape, RoHS Compliant, Zetex Inc, ZXMP6A17E6TA (Digi-Key p/n ZXMP6A17E6CT-ND $1.04/1) http://www.zetex.com/3.0/pdf/ZXMP6A17E6.pdf.

Purpose:

• This is the external PMOSFET of U250 which will turn off given that there is one or more of the four trigger events as described earlier. See U250. U250 uses the RDS(on) of the saturated Q250 as a current sense resistor which generates a Vds voltage which is detected across the Vs (Pin 1) and DRAIN (Pin 2) pins and if it is greater than some threshold voltage, U250 will switch Q250 off thus breaking the circuit.

Specifications/ Calculations:

• The maximum SPS output current specified was 400 mA. There are three circuit-breaker threshold voltage versions of U250: 300 mV, 400 mV and 500 mV. For certain reasons the 300 mV threshold part was chosen. See U250. Therefore the R_DS(on) of the PMOS should be around 300 mV / 400 mA = 0.75 ohm. We used a value of 1 ohm. See R252. Even though the PMOS is used as a switch (cutoff and saturation) and not an amplifier (cutoff, triode and saturation) we wanted to remove dependence of U250's threshold voltage detection from the less precise R_DS(on) of Q250 and to a more precise sense resistor. Therefore we added a current sense resistor (R252) in series with the drain of Q250 to produce a circuit-breaker resistor, Rcb, which is the series combination of Q250's R_DS(on) and R252 between the two pins 1 and 2 of U250. We chose a PMOS with a low R_DS(on) compared to the needed calculated value needed to trip the circuit-breaker thereby making R252 close to Rcb in value. Therefore Q250 is used mostly as a switch and the voltage drop across R252 is used to trigger the switch. See R252. The breakdown voltage of Q250 has to be greater than 20 V and should have a low "turn on" capacitance. We do not care so much about the Vt but it does affect the "turn on" capacitance but these factors were not considered.

MAX5902AAETT Circuit-Breaker External P-MOSFET RDS(on) Additional Series Resistor (R252)

Part Description:

• 0.82 ohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Panasonic - ECG, ERJ-6RQFR82V (Digi-Key p/n P.82DCT-ND $2.10/10) http://www.panasonic.com/industrial/components/pdf/AOA0000CE3.pdf.

Purpose:

• This resistor dominates the circuit-breaker resistor's (Rcb) value. It is in series with the drain (hence RDS_(on) of Q250 to make up Rcb. The voltage drop across it is used to detect an overcurrent event given that it is greater than 300 mV. See Q250 and U250. Note: This value may change due to board level testing results.

Specifications/ Calculations:

• Since the typical value of R_DS(on) of Q250 is 0.125 ohm and Rcb was about equal to 1 ohm, R252 = Rcb - R_DS(on) = 0.875 ohm. The closet standard value was 0.82 ohm. Given this value of Rcb and the circuit-breaker trip threshold voltage of 300 mV, the maximum SPS current which can be drawn before an overcurrent event is I_max = 300 mV / (0.125 ohm + 0.82 ohm) = 317 mA which is under the specified maximum SPS current spec.

Overvoltage Protection

Undervoltage/Overvoltage Protection (U251)

Part Description:

• Nanopower Push-Pull Output Comparator with Voltage Reference, 1.8 V < Vin < 5.5 V, SOT-23-6, Tape & Reel (TR), RoHS Compliant, Texas Instruments, TLV3012AIDBVT (Digi-Key p/n 296-16830-2-ND $262.50/250) NOTE: Ordered samples from vendor no Digi-Key. http://focus.ti.com/lit/ds/symlink/tlv3012.pdf.

Purpose:

• This comparator compares the specified divided SPS output voltage (see R254, R255) to its internal reference voltage (1.242 V) for an overvoltage trigger event at the SPS output. It is powered by a secondary supply consisting of CR250 and C250. Also it has a pseudo low-pass filter consisting of C251 and its output (pin 1) with the input being pin IN+ (pin 3). See CR250, C250 and C251 respectively.

Specifications/ Calculations:

• We wanted a low power push-pull output comparator to get rail to rail output swing (approximately 200 mV to 3.1 V) and have reasonable switching and rise/fall times, on the order of several microseconds and nanoseconds respectively. We tied the IN- pin (pin 4) to the internal reference voltage REF pin (pin 5) which will be compared to the divided SPS output voltage at its IN+ pin (pin 3). See R254, R255 and C251. The "undervoltage" protection is actually provided by CR250 and C250 where U251 will remain powered for a specified amount of time if the +3.3 V SPS output rail drops. See CR250 and C250.

TLV3012AIDBVT Overvoltage Detection External N-MOSFET Current Limiting Resistor (R253)

Part Description:

• 47.0 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF4702 (Digi-Key p/n RHM47.0KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R253 along with Q251 will "emulate" an overcurrent trigger event as seen by U250 when an overvoltage at the SPS output trigger event as seen by U251 occurs. See Q251. When the SPS +3.3 V output rises above a certain threshold, the output of U251 goes high, turning Q251 on. When this happens it pulls pin 2 of U250 very close to ground and current flows through R253 and Q251. Now that pin 2 is close to ground and pin 1 is normally close to the power bus voltage this is much greater than 300 mV this causing an overcurrent trigger event for U250. R253 limits the extra current pulled through Q251 when it is turned on.

Specifications/ Calculations:

• In normal SPS operation, the voltage drop across Rcb will be less than 300 mV and R253 is connected to the high impedance pin 2 of U250 so no current flows through it. If there is an overvoltage trigger event at the SPS output, Q251 is turned on conducting current through R253 which will have a voltage drop of approximately 300 mV less than the power bus voltage: V_R253 = 16.8 V - 300 mV = 16.5 V. This results in a current boost of about I_R253 = 16.5 V / 47 kohm = 351 uA which is negligible.

TLV3012AIDBVT External Logic-Level N-MOSFET (Q251)

Part Description:

• 100 V, 170 mA, RDS(on) = 10 ohm @ Vgs = 4.5V, SOT-23, Cut Tape, RoHS Compliant???, N-Channel Logic-Level MOSFET, Infineon Technologies, BSS123E6327 (Digi-Key p/n BSS123INCT-ND $0.36/1) http://rocky.digikey.com/WebLib/Infineon/Web%20Data/BSS123.pdf.

Purpose:

• This is a logic-level NMOSFET. When an overvoltage at the +3.3 V SPS output occurs, U251 will output a logic high turning Q251 on and thus conducting current through R253. The current flowing through R253 also flows through Rcb and its magnitude is dependent on the value of R253 and the bus voltage at the time (nominal value of 16.8 V). When there is no overvoltage at the +3.3 V SPS output U251 outputs a logic low thus keeping Q251 off.

Specifications/ Calculations:

• Since the gate of this FET would be driven by the output of a comparator in U251 it would be best for the FET to be a logic-level device. Other concerns was for the drain-source breakdown voltage to be higher than 30 V as the highest possible DC value the bus voltage rail would be is 20 V.

TLV3012AIDBVT UVLO Lockout Resistor Network (R254, R255)

R254

Part Description:

• 18.2 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF1822 (Digi-Key p/n RHM18.2KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R254 along with R255 form a voltage divider with respect to the +3.3 V SPS output rail. When there is an overvoltage at the +3.3 V SPS output the voltage at the IN- pin (pin 4) of U251 will be greater than the internal reference voltage of U251 (typically 1.242 V) and will result in the comparator in U251 outputing a logic high value. When the SPS output is below a certain threshold the input voltage (pin 4) to U251 is less than the internal reference voltage and the comparator's output is a logic low.

Specifications/ Calculations:

• From the LPC2148 datasheet the maximum supply voltage it can handle is 3.6 V therefore we specified that if the +3.3 V SPS output was to reach 3.5 V we would want this to qualify as an overvoltage trigger event. Since we have been using several 10.0 kohm resistors we specified R255 to be 10.0 kohm. Therefore using the overvoltage trigger event value to be 3.5 V and the compared voltage to be 1.242 V we solved for R254: 1.242 V = (3.5 V * R255)/ (R255 + R254) and solving for R254 = 18.18 kohm. The closet standard value was 18.2 kohm.

R255

Part Description:

• 10.0 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF1002 (Digi-Key p/n RHM10.0KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R255 along with R254 form a voltage divider with respect to the +3.3 V SPS output rail. When there is an overvoltage at the +3.3 V SPS output the voltage at the IN- pin (pin 4) of U251 will be greater than the internal reference voltage of U251 (typically 1.242 V) and will result in the comparator in U251 outputing a logic high value. When the SPS output is below a certain threshold the input voltage (pin 4) to U251 is less than the internal reference voltage and the comparator's output is a logic low.

Specifications/ Calculations:

• We specified R255 = 10.0 kohm. See R254.

TLV3012AIDBVT "Secondary Power Supply" Schottky Diode (CR250)

Part Description:

• 30 V, 1.5 A, 4 ns, New MiniPower 2P, Cut Tape, RoHS Compliant???, Panasonic - SSG, MA2Q70500L (Digi-Key p/n MA2Q70500LCT-ND $0.83/1) http://www.semicon.panasonic.co.jp/ds/eng/SKH00017BED.pdf.

Purpose:

• CR250 and C250 form U251's power supply. This diode prevents C250 from discharging anywhere but to the V+ supply pin of U251. U251 is indirectly powered by the +3.3 V SPS rail. Pin 6 (V+) of U251 will be charged to a value very close to the +3.3 V SPS rail. As U251 draws more current when needed and its V+ voltage drops CR250's Vf below the SPS voltage CR250 will re-charge C250. Therefore the average DC current through CR250 is not easily calculable but will be on the order of tens to hundreds of uA. Schottky diodes were chosen for their fast switching and reverse recovery times. In response to a overvoltage event at the +3.3 V SPS output, U251 will output a logic high and turn Q251 on which will cause an overcurrent event at U250 which will in response turn Q250 off thus circuit breaking the bus rail from the SPS and the +3.3 V SPS output voltage rail will go to zero. That is the sequence of events without delay times. This is the way U251 emulates an overcurrent event from an overvoltage event.

Specifications/ Calculations:

• CR250 has a low forward voltage and U251 has an input voltage supply range of 1.8 V to 5.0 V so when the SPS voltage is being brought up C250 is being charged through CR250 leaving the voltage of SPS minus the Vf of CR250 at pin 6 (V+) of U251: V+ = 3.3 V - 0.05 V = 3.25 V (approximately). See the first graph on page 2 of the MA2Q705 datasheet given that the steady state nominal forward current through CR250 < 1 mA. See C250.

TLV3012AIDBVT "Secondary Power Supply" Cap (C250)

Part Description:

• 2.7 uF, 10 V, 0805, X5R, Cut Tape, RoHS Compliant, Kemet, C0805C275K8PACTU (Digi-Key p/n 399-3127-1-ND $7.02/10) http://www.kemet.com/kemet/web/homepage/kechome.nsf/vapubfiles/F3102X5R.pdf/$file/F3102X5R.pdf.

Purpose:

• Along with CR250, C250 forms the power supply for U251. C250 is charged to a voltage less a forward diode drop (see CR250) from the +3.3 V SPS output rail under normal operating conditions. U251 draws a constant 2.8 uA supply current so CR250 is always trickle charging C250, therefore the voltage across C250 will be Vf_CR250 = 0.37 V (this is worst case Vf) less than 3.3 V. See CR250. However U251 can operate from 1.8 V to 5.0 V. Since the voltage at V+ of U251 is approximately 3.0 V and the lower limit of the supply voltage range of U251 is 1.8 V, C250 has to be able to store enough charge such that if the SPS output drops down by a certain amount of voltage, U251 is still powered for a certain amount of time thus preventing U251 from power cycling if the SPS output ramps back up to +3.3 V. We want to prevent this because as U251 is powering up the comparator could possibly switch. That behavior has to be observed in experiment but we assume that the initial state of the comparator will be logic low. If the SPS output toggles or drops in value we want U251 to have power for a specified amount of time in case the magnitude of the voltage change of a transient causes a trigger event but lasts a very short amount of time or would normally shut down U251, thus avoiding the time needed for U251 to power cycle. When the overvoltage at the SPS output event occurs there is a finite amount of time required before Q250 is eventually turned off namely the propagation delay of the comparator in U251, the turn-on delay time of Q251, the time to turn off Q250 by U250 and the turn-off delay time of Q250. These typical times as are 12 us, 8 ns, 4 us and 26.2 ns as specified on pages 3, 3, 8 and 4 in the TLV3012, BSS1223, MAX5902 and ZXMP6A17E6 datasheets respectively, resulting in an ideal propagation delay of about 17 us. After these propagation delay times, Q250 is off and the input voltage Vin (pin 2) of U200 is zero volts (after C201 is discharged) triggering the SHDN\ pin and turning off U200. When this happens the power supply to U251 is essentially removed, so the time constant for C250 has to be long enough such that as U250 is going through its start sequence (150 ms) the comparator in U251 can output a logic low and turn Q251 off as the voltage it is comparing, IN+ (pin 3), to its internal reference voltage, REF, is the divided (see R254, R255) SPS output rail voltage which at this time is zero volts. To make sure that this sequence of events happen we specified U251 to have power long enough to turn Q251 back off while U250 is turning back on again. Under normal operating conditions (no UVLO event) and assuming that there are no long or catastrophic transients, U251 should always be on. This design will always keep U250 on which will minimize any unknown states at the comparator output resulting from U251 turning off, then on again, etc. Upon initial power up we assume that the output of U251 will be logic low keeping Q251 off to prevent a false overcurrent event for U250 which may prevent the SPS from working as U250 will never turn Q250 on and will just cycle. This seems unlikely because upon initial power up, U251 has no power and cannot output logic high. However as the +3.3 V SPS output is being brought up the output of the comparator is undefined which is not good being directly connected to the gate of Q251 but we still think that Q251 will remain off or will rapidly switch off if it is ever on after the transients.

Specifications/ Calculations:

• We calculated the needed amount of charge, Qt, C250 would have to store such that U251 would have power for at least 0.5 s (our specified amount of time U251 should have power during these events) given that the SPS output voltage dropped by 1 V from which we calculated C250's capacitance. C250 = [((Qt * 1.2) + (Iq * tp))] / Vp, where Qt = [((Input Capacitance) * Vt) + ((Reverse Transfer Capacitance) * Vin)], Iq = 2.8 uA, ts = 0.5 s, Vp = 1 V Qt is the sum of products of the input capacitance of Q251 times the maximum threshold voltage plus the reverse transfer capacitance of Q251 times the maximum Vds swing, namely Vin. Iq is the supply current of U251, tp is the amount of time we want U251 to have power, Vp is the amount of voltage the SPS output drops and the 1.2 term is a fudge factor because Qt is dependent on some other factors not explicitly shown. tp was specified to be 0.5 s, this time is the time C250 can supply power to U251 which is longer than the propagation delays mentioned above including some margin just in case any trigger events do not go away and U250 has to go through another 150 ms start sequence. If the trigger events remain longer than the 0.5 s, then U251 turns off and the whole SPS will go through another initial power up sequence. Vp = 1 V, i.e. the input voltage V+ (pin 6) to U251 can drop to about 3.0 V - 1 V = 2 V. Iq = 2.8 uA, see page 3 in the TLV3012 datasheet. Qt = (85 pF * 2 V) + (15 pF * 20V) = 470 pC, see page 3 in the BSS123 datasheet. C250 = 1.40056 uF we decided to chose a 2.7 uF cap to give us a little more tp due to any unknown delays and the like we did not consider.

TLV3012AIDBVT Overvoltage Detection Cap (C251)

Part Description:

• 0.015 uF, 100 V, 0805, X7R, Cut Tape, RoHS Compliant, AVX Corporation, 08051C153KAT2A (Digi-Key p/n 478-1359-1-ND $2.64/10) http://rocky.digikey.com/WebLib/AVX/Web%20Data/X7R%20(C).pdf.

Purpose:

• C251 is used as a low-pass filter to node IN+ (pin 3) of U251 and as positive feedback to make the comparator switch faster and to make sure that once the comparator is switching it completes the transistion. Under normal SPS operation, when U251 is keeping Q251 off, the OUTPUT (pin 1) of U251 is at zero volts thus the cap is acting like a low-pass filter, the node connected to pin 3 of U251 is the input to the filter. If there are transients at the +3.3 V SPS output this node will also experience proportional transients. If the magnitude of these transients are great enough (but fast in duration) then U251's comparator switches, which is undesirable so we want true ovevoltage events to trigger the comparator. C251 will remove most of these false event transients. When there is a true overvoltage event the comparator starts to switch. If there is another transient (false event) where the magnitude of the voltage goes below the threshold the comparator could possibly try to switch back. We want the comparator to avoid reacting to false transients. C251 prevents this because as the comparator is rising its output voltage, C251 raises the voltage on pin 3 of U251 thus reinforcing the comparator to keep on raising its output voltage. This is positive feedback. Also C251 decreases the rise time of the comparator. Basically if there is something weird going on at the SPS output, i.e. it is oscillating between 0 V and 3.3 V, C251 will help to make sure that U251 turns Q251 off, eventually turning off Q250 which will give a 150 ms time period for the weird things to go away, given that U251 does not shutdown during these transients. See C250.

Specifications/ Calculations:

• We wanted C251 to filter transients which lasted less than 100 us therefore we need to find the output resistance C251 sees under a transient (or AC) condition. We used the zero-time coefficient technique to solve for the resistance and eventual capacitance. Under a transient condition the SPS output and comparator output are grounded (DC voltage) and removing C251 the resistance it sees is the parallel combination of R254 and R255. R254 || R255 = 6.46 kohm, with an RC = 100 us we can solve for C = 15.4 nF. The closest standard value is 0.015 uF. The voltage rating needs to be greater than 20 V just for safe measure.

Switcher

Step-Down Switching Voltage Regulator (Buck Topology) (U200)

Part Description:

• Main SPS Switching Buck Regulator at +3.3V rail and Io,max ~ 400 mA. Monolithic Step-Down Buck Switching Regulator, Current Mode Control, 1.5 A, 1.25 MHz (Adjustable), 3V < Vin < 25 V, 8-MSOP (No exposed ground/thermal pad), RoHS Non-Compliant, Linear Technology, LT1767EMS8 (Digi-Key p/n LT1767EMS8-ND $6.00/1) http://www.linear.com/pc/downloadDocument.do?navId=H0,C1,C1003,C1042,C1032,C1064,P1915,D1885.

Purpose:

• This is the switching voltage regulator. It specifically is a buck topology which uses current controlled loop control and has adjustable voltage at the output.

Specifications/ Calculations:

• We wanted a switcher which had an internal switching frequency of at least 1 MHz, low switch resistance, could supply at least 1 A of current, had a shutdown or disable pin, could be synchronized with an external clock and was in a small package. We borrowed U200 from the previous LV2 SPS design because for the given PSAS specs it was the best fit. There is no "direct" output undervoltage protection within the SPS, however if the voltage at the output starts to fall due to an increasing load the circuit-breaker protection will kick in. See U250. There is a microcontroller supervisory circuit (U283) in the GLUE logic section which will reset U280 if the +3.3 V rail drops below a certain threshold voltage (3.075 V version) for a specified amount of time. However U283 cannot correct the fault condition but only keeps U280 reset given that the SPS output rail is still below the threshold voltage. Other than an overcurrent event at the output the quality of regulation by the SPS (specifically U200) will dictate if the SPS output rail drops. See the GLUE Logic section for clock synchronization with X281 and U281 and U200's SYNC pin (pin 8).

LT1767EMS8 UVLO Lockout Resistor Network: (R209, R210)

R209

Part Description:

• 60.4 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF6042 (Digi-Key p/n RHM60.4KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R209 is part of the UVLO resistor divider for U200. Along with R210 this resistor divider ensures that U200 does not regulate until after a turn-on threshold voltage level has been met.

Specifications/ Calculations:

• The UVLO voltage was specified to be 9 V. See page 10 and Figure 4 in the LT1767 datasheet. Letting R210 = 10.0 kohm and VH = 9 V, the UVLO formula from page 10 in the datasheet is R210 = 1.33 V / ((VH - 1.33 V)/R209 - 3 uA) and solving for R209 = 59 kohm. The closet standard value was 60.4 kohm.

R210

Part Description:

• 10.0 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF1002 (Digi-Key p/n RHM10.0KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• This is a part of the UVLO resistor divider of the LT1767EMS8 (U200). Along with R210 this resistor divider ensures that the LT1767EMS8 does not regulate until after a turn-on threshold voltage level has been met.

Specifications/ Calculations:

• The UVLO voltage was specified to be 9 V. See page 10 and Figure 4 in the LT1767 datasheet. Letting R210 = 10.0 kohm and VH = 9 V, the UVLO formula from page 10 in the datasheet is R210 = 1.33 V / ((VH - 1.33 V)/R209 - 3 uA) and solving for R209 = 59 kohm. The closet standard value was 60.4 kohm.

LT1767 UVLO Resistor Network Shunt Cap (C210)

Part Description:

• 0.1 uF, 50 V, 0805, X7R, Cut Tape, RoHS Compliant???, BC Components , VJ0805Y104KXATW1BC (Digi-Key p/n BC1298CT-ND $0.72/10) http://rocky.digikey.com/WebLib/BC%20Components/Web%20Data/MLCC,%20SMT%20NPO%20(10,16,25%20%26%2050V).pdf.

Purpose:

• This cap is a noise bypass cap to the SHDN\ pin of U200.

Specifications/ Calculations:

• We specified a 0.1 uF (standard value) bypass cap.

Buck Input Cap (C201)

Part Description:

• 10 uF, 25 V, 1206, X5R, Cut Tape, RoHS Compliant, Panasonic-ECG, ECJ-3YB1E106K (Digi-Key p/n PCC2414CT-ND $5.45/10) http://www.panasonic.com/industrial/components/pdf/abj0000ce4.pdf.

Purpose:

• C201 is the input cap to U200.

Specifications/ Calculations:

• From page 7 in the LT1767 datasheet we used the formula to calculate the rms ripple input current: Irms = Io * sqrt(Vout * (Vin - Vout) / (Vin)^2). Given that the unlikely (transient) worst case power bus supply current drawn by the SPS is Io = 1 A, Vin = 20 V and Vout = 3.3 V, we can solve for Irms = 1 A * sqrt(3.3 V * (20 V - 3.3 V) / (20 V)^2) = 371 mA. The ripple voltage is equal to: dV = (Irms * dt) / C201, where T = 1/f = 667 ns. The ripple voltage at the input is not too critical given that it is much less than a few volts in magnitude which could trigger the SHDN\ pin of U200. Other than that the only other problem is that if the current through L201a,b is constant and there is a sudden step in the current draw the voltage across C201 could possibly become greater than the power bus voltage. There even is potential of ripple voltage actually aiding U200 efficiency where the ripple voltage could possibly increase the duty cycle of U200. Given the reasoning’s above the ripple voltage is not too much of a concern. Therefore we just want a bigger cap in size (1206) and value such that it can act as a temporary voltage supply to U200 under slightly larger current loading. We chose to use a 10 uF cap in a 1206 package. The ripple voltage then is: dV = (Irms * dt) / C201 = (371 mA * 1/f) / 10 uF = 24.7 mV. This is negligible. Since this is a high frequency node C201 and CR200 will be closely grounded together.

LT1767 Frequency Compensation Caps (C205, C206)

C205

Part Description:

• TBD

Purpose:

• Part of the frequency compensation of U200.

Specifications/ Calculations:

• See pages 24 - 25 of the CAN Node Switch Mode Power Supply (SPS) (200) in the Component Design for LV2 Power Electronics (Except Main Battery) engineering design notes. Also see pages 48 - 50 of the Linear Technology application note an-19 and see application note an-76. The value is TBD.

C205 may not be needed given that the frequency compensation of R206/C206 is satisfactory.

C206

Part Description:

• TBD

Purpose:

• Part of the frequency compensation of U200.

Specifications/ Calculations:

• See pages 24 - 25 of the CAN Node Switch Mode Power Supply (SPS) (200) in the Component Design for LV2 Power Electronics (Except Main Battery) engineering design notes. Also see pages 48 - 50 of the Linear Technology application note an-19 and see application note an-76. The value is TBD. To find the value of R206/C206 Linear Technology suggested that this was a trail and error experimental process. An experimental test circuit would vary the load at the output and the transient response of the output would be recorded, namely the voltage ripple at the output. For example, in an experiment, R206 is held constant and C206 is swept until a desirable output transient response is observed or vice versa where C206 is held constant and R206 is swept and then the constant parameter is stepped and another experimental sweep of the variable parameter is repeated. The standard starting value for C206 is about 1 nF and is usually decremented.

Boost Cap (C202)

Part Description:

• 0.1 uF, 50 V, 0805, X7R, Cut Tape, RoHS Compliant, Panasonic-ECG, ECJ-2YB1H104K (Digi-Key p/n PCC1840CT-ND $1.61/10) http://industrial.panasonic.com/www-data/pdf/ABJ0000/ABJ0000CE1.pdf.

Purpose:

• Boost cap C202 is connected to the BOOST pin (pin 1) on U200. It is used to step up the voltage from Vsw (pin 3) of U200 to drive the internal switch.

Specifications/ Calculations:

• The LT1767 datasheet recommends using a 0.1 uF film or ceramic cap with an ESR < 1 ohm (see page 9). The ECJ-2YB1H104K datasheet does not specify the ESR but states that this family of caps have a low ESL. Also this same family of caps were used in the previous LV2 SPS.

Boost Rectifier Diode (CR201)

Part Description:

• 180 V, 0.6 A, SOT-23, Cut Tape, RoHS Compliant, Micro Commercial Co., MMBD1501-TP (Digi-Key p/n MMBD1501TPMSCT-ND $2.00/10) http://59.120.39.77/mccsemi/up_pdf/MMBD1501(A)-1505(A)(SOT-23).pdf.

Purpose:

• This diode is used to charge the boost cap C202.

Specifications/ Calculations:

• It should have a voltage rating >> 20 V and a current rating of several hundred mA. Since this is a high power diode in the SPS we wanted the package to be bigger than the MiniPower 2P but not to big, so we chose an SOT-23.

Buck (Catch) Schottky Diode (CR200)

Part Description:

• 30 V, 1.5 A, 4 ns, New MiniPower 2P, Cut Tape, RoHS Compliant???, Panasonic - SSG, MA2Q70500L (Digi-Key p/n MA2Q70500LCT-ND $0.83/1) http://www.semicon.panasonic.co.jp/ds/eng/SKH00017BED.pdf.

Purpose:

• This is the buck (catch) output diode.

Specifications/ Calculations:

• Using the formula on page 9 in the LT1767 datasheet we calculated the average DC current that CR200 should be able to handle. I_Davg = Io (Vin - Vout) / Vin, where Io is the output current of the SPS, Vin is the voltage at the input (pin 2) of U200 and Vout is the SPS output voltage. With Vout = 3.3 V and using the worst case Io = 1 A and Vin = 20 V values Id,avg = 835 mA. Even though the maximum specified Io = 400 mA and F200 is rated at 500 mA, we used Io = 1 A because the fuse had a finite opening time of about 1 s upon which L200a,b could draw higher currents through CR200 (not through U250) under certain events. Also the unlikely maximum bus voltage is 20 V, which we should never see, but which is worst case. With Id,avg = 835 mA, we want CR200 to be rated at a higher forward current. Also the reverse voltage needs to be greater than 20 V and the forward voltage needs to be low. We chose a part with: If = 1.5 A, Vr = 30 V and Vf < 0.37 V. The forward current, If, spec is a little overkill but since it is in a small 2 pin package we like it. Since this is a high frequency node CR200 and C201 will be closely grounded together.

Split Buck Inductor (L200a, L200b)

Part Description:

• SD3118-470-R Low Profile Power Inductor 47 uH, Shielded Drum Package (Bobbin), RoHS Compliant???, Tape and Reel. http://www.cooperet.com/library/products/PM-4129%20SD3118%20Series.pdf.

Purpose:

• This is the main buck output inductor. L200a and L200b are one inductor but are split such that we can have a secondary output voltage supply of 5 V (see CR251 and C252). We chose the shielded-drum or bobbin style of package because they are smaller than similar torroid packages.

Specifications/ Calculations:

• For a worst case scenario we expect to run the inductor's at an ambient temperature of 80 degrees C. That is, the SPS is off and the inductors are not dissipating any power. We also want the inductor's saturation current to be larger than 400 mA and be big enough both in value and physical size such that they will not easily go into discontinuous mode or burn up due to power/heat dissipation. From page 2 in the SD3118-470-R datasheet with a 40 degree C rise in temperature (i.e. running at 120 degree C) the total power loss due to the inductor is 200 mW. Summing the non-linear core losses and DC power losses should be less than this total power loss. We went through a trial and error process of finding the inductor value given that the saturation current should be greater than 400 mA and the range of current that the inductor will carry is about 10 mA to 300 mA. We specified the average DC current being equal to 300 mA. After using the formulas and graphs in the SD3118-470-R datasheet we found that we wanted an inductor of approximately 100 uH. Since the inductor is split we had to use two 47 uH inductors. Given f = 1.5 MHz, L200a = 47 uH, Vin = 20 V, Vout = 3.3 V, Idc = 300 mA, Vd = 0.37 V (see CR200 datasheet) and Vsw = Rsw * Idc = 66 mV (see U200 datasheet for Rsw) and K = 12 and DCR = 1.21 ohm (taken from the SD3118-470-R datasheet). From the table on page 1 in SD3118-470-R datasheet the peak-to-peak magnetic field is given by the formula in note (4): Bp-p = K * L200a * delta_I, where K is taken from the table, Bp-p and L200a are already in units of mT and uH respectively. To find the ripple inductor current, delta_I, we calculate the applied volts-microseconds across L200a,b and find delta_I from the inductor current differential equation: V = L200a * di/dt. The applied volts-microseconds (V * dt) can be found using the following calculations: V * dt = (vin - Vout) * T * D, where T is the period of the switching frequency (1/f) and D is the duty cycle of U200. Using the formula on pages 1 and 2 in the National Semiconductor Application Note An-1197: D = (Vout + Vd) / (Vin + Vd - Vsw). Therefore D = (3.3 V + 0.37 V) / (20 V + 0.37 V - 66 mV) = 18.1% T = 1/f = 1/(1.5 MHZ) = 667 ns The applied volts-microseconds therefore equals to V * dt = (20 V - 3.3 V) * 667 ns * 0.181 = 2.01 V * us Given L200a = 47 uH we can solve for di = delta_I = (V * dt) / L200a = 42.82 mA Since the applied volts-us we calculated is across the whole 100 uH (47 uH * 2 = 94 uH) inductance we need to cut this value in half since the all the parameters taken from the SD3118-470-R datasheet is for a single 47 uH inductor. Therefore the di we will be using for further calculations is delta_I' = di' = di/2 = 21.4 mA. We need to solve for Bp-p to find the core losses from the graph on page 3 in the SD3118-470-R datasheet. Bp-p = K * L200a * di' = 12 * 47 H * 21.4 mA = 12.1 mT The graph on page 3 of the SD3118-470-R datasheet does not include a switching frequency of 1.5 MHz but estimating from the spacing between the different frequencies we extrapolated the core losses to be equal to about 11 mW. The DC power losses is simply the direct-current resistance (DCR) times the average DC current which is Pdc = (Idc)^{2 * DCR = (300 mA)}2 * 1.21 ohm = 109 mW. The DC power loss and non-linear core losses sum to 109 mW + 11 mW = 120 mW < 200 mW so this inductor should work.

Buck Output Cap (C200)

Part Description:

• 22 uF, 6.3 V, 0805, X5R, Cut Tape, RoHS Compliant, Panasonic - ECG , ECJ-2FB0J226M (Digi-Key p/n PCC2401CT-ND $8.31/10) http://dkc3.digikey.com/PDF/T062/1235.pdf.

Purpose:

• This is the buck output switching regulator capacitor. To maintain more precise regulation the capacitance should be constant as much as possible therefore a X5R, X7R or NPO temperature coefficient cap should be used.

Specifications/ Calculations:

• Since this is the buck output cap and there is overvoltage protection at the output (see U251), it only needs a voltage rating a little greater than 3.3 V. We found a 6.3 V rated part. We would prefer to have a smaller cap than say C203 so a ceramic part was used. Using the formula on page 23 in the Linear Technology Application Note AP19: C200 = [1 / (8 * (L200a+L200b) * f^2)] / [Vpp / (Vout * (1 - Vout/Vin))], where (L200a + L200b) is the 100 uH (47 uH * 2 = 94 uH) buck output inductor, Vout = 3.3 V, Vin = 20 V, f = 1.5 MHz, Vpp is the output voltage ripple and the ESR term is dropped. In contrast to C201 the output voltage ripple is significant because we want an ideal +3.3 V DC output voltage. The maximum output voltage ripple was specified to be Vpp = 5 mV. The previous LV2 SPS design used a 22 uF buck output cap and we wanted to use the same value. Using the formula to solve for the buck output inductor it equated to about 1.4 uH. We would want to use that value of inductance if we needed to supply higher output currents as the smaller inductor would have a higher saturation current, lower DC power losses but larger core losses. The biggest concern however in not using the smaller inductor was staying away from discontinuous mode operation. We did not want to do this so we ignored the dependence of C200 to L200a,b as is shown by the formula and found L200a,b as wee did. See (L200a, L200b). Using the formula again with the 100 uH (94 uH) inductor and solving for Vpp we got, Vpp = 74 uV, which is negligible. Under nominal operating conditions, we expect a flat 3.3 V output.

LT1767 FB Resistor Network: (R200, R201)

R200

Part Description:

• 10.0 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF1002 (Digi-Key p/n RHM10.0KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• This along with R201 forms the feedback (FB) resistor network which sets the value of Vout of U200.

Specifications/ Calculations:

• The LT1767 datasheet on page 7 suggested R200 to be 10 kohm.

R201

Part Description:

• 17.4 kohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant???, Rohm, MCR10EZHF1742 (Digi-Key p/n RHM17.4KCCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• This along with R200 forms the feedback (FB) resistor network which sets the value of Vout of U200.

Specifications/ Calculations:

• Using the formula on page 7 in the LT1767 datasheet: R201 = R200 * (Vout - 1.2 V) / (1.2 V - R200 * 0.25 uA), where R200 = 10.0 kohm and Vout = 3.3 V R201 equated to 17.54 kohm the closet standard value was 17.4 kohm.

LT1767 FB Resistor Network Shunt Caps (C208, C209)

C208

Part Description:

• TBD

Purpose:

• Part of the frequency compensation of U200.

Specifications/ Calculations:

• See pages 24 - 25 of the CAN Node Switch Mode Power Supply (SPS) (200) in the Component Design for LV2 Power Electronics (Except Main Battery) engineering design notes. Also see pages 48 - 50 of the Linear Technology application note an-19 and see application note an-76. The value is TBD.

C209

Part Description:

• 10 pF, 50 V, 0805, NPO, Cut Tape, RoHS Compliant???, BC Components, VJ0805A100JXACW1BC (Digi-Key p/n BC1256CT-ND $0.52/10) http://www.cooperet.com/library/products/PM-4313%20CMS-Series.pdf.

Purpose:

• Part of the frequency compensation of U200.

Specifications/ Calculations:

• We used the same value as from the LV2 design. See pages 24 - 25 of the CAN Node Switch Mode Power Supply (SPS) (200) in the Component Design for LV2 Power Electronics (Except Main Battery) engineering design notes. Also see pages 48 - 50 of the Linear Technology application note an-19 and see application note an-76.

SPS Output Power On LED Current Limiter Resistor (R214)

Part Description:

• 649 ohm, 0805, 1%, 1/8 W, Cut Tape, RoHS Compliant, Rohm, MCR10EZHF6490 (Digi-Key p/n RHM649CCT-ND $0.38/10) http://www.rohm.com/products/databook/r/pdf/mcr10.pdf.

Purpose:

• R214 is the SPS Output Power On LED current limiting resistor.

Specifications/ Calculations:

• From the datasheet Vf = 1.9 V, resulting in a current limiting resistor of about, R214 = (3.3 V - 1.9 V) / (2 mA) = 650 ohms. The closest standard value was 649 ohms.

Secondary Voltage Supply

SPS Secondary Buck (Catch) Schottky Diode (CR251)

Part Description:

• 30 V, 1.5 A, 4 ns, New MiniPower 2P, Cut Tape, RoHS Compliant???, Panasonic - SSG, MA2Q70500L (Digi-Key p/n MA2Q70500LCT-ND $0.83/1) http://www.semicon.panasonic.co.jp/ds/eng/SKH00017BED.pdf.

Purpose:

• CR251 along with C252 and L200a form the second buck switching voltage power regulator which will be eventually regulated down to 5 V possibly with a low-dropout (LDO) linear voltage regulator. For consistency we used the same Schottky diode as CR200. Also the general understanding was that this secondary buck will power specific parts like 5 V ADCs on certain nodes (like the IMU) and we expect that this diode's rated specs are more than enough.

Specifications/ Calculations:

• Using the formula on page 9 in the LT1767 datasheet we calculated the average DC current that CR251 should be able to handle. Id,avg = Io (Vin - Vout) / Vin, where Io is the secondary output current of the SPS, Vin is the voltage at the node which is between L200a and L200b and Vout is the SPS secondary output voltage. We assumed that Vin would be switching somewhere between 1.6 V and 7 V. With Vout = 5 V and using the worst case Io = 1 A and Vin = 7 V values Id,avg = 286 mA (again this diode is overrated). We knew that not every node would need a secondary 5 V supply but even the ones that did, the added current should not cause overcurrent events (see U250). Hopefully.

SPS Secondary Buck Output Cap (C252, C252a)

Part Description:

• TBD

Purpose:

• This cap along with CR251 and L200a form the second buck switching voltage power supply which will be eventually regulated down to 5 V possibly with a low-dropout (LDO) linear voltage regulator. These are application specific caps whose values are mostly independent from the SPS design.

Specifications/ Calculations:

• The only difference between C252 and C252a are the packages and that only one of them will actually be on the PCB. Since we do not know any details about the actual application specific circuitry each SPS will power from an SPS design point of view, we chose to use both a 0805 and 1206 package. We chose two packages because we moved all relevant parts to the 0805 package from 1206 as in the LV2 SPS design and in case a specific application node needs a more beefy cap a 1206 package cap can be used. The lay out of the parts will not be side by side as suggested in the schematic but are offset and superimposed on top of each other on the same side of the PCB. Because only one cap will be used we offset the pads such that they are not directly on top of each other and either package can be placed down thus saving space. The values are TBD.

Power LED

SPS Output Power On LED (D201)

Part Description:

• 1.9 V, 90 mcd @ 20 mA, 609 nm, 0805, Orange Diffused LED, CML Innovative Technologies Inc, CMDA5BA7D1S (Digi-Key p/n L71515CT-ND $3.00/10) http://www.chml.com/pdf/temp/CMDA5BA7D1S.pdf.

Purpose:

• This is an orange LED which is lit given that the nominal SPS 3.3 V rail is up. It is mainly used as an initial indicator of the 3.3 V rail's status. Orange was an arbitrary choice, however any other LEDs in the Glue Logic section needed to be different colors. The intensity and viewing angle are not critical since the only time the information from the LED is useful is in trouble shooting on the ground. It remains lit throughout the whole flight.

Specifications/ Calculations:

• From the CMDA5BA7D1S datasheet, Vf = 1.9 V. We specified the LED drive current to be 2 mA. See R214 for I-V calculations.

Bill of Materials (BOM)

NOTE: Component fields with a "-" mean they are TBD.

Digi-Key BOM

Coiltronics BOM